The usage of light-emitting diodes (LEDs) to provide illumination is increasing rapidly as the cost of LEDs decrease and the endurance of the LEDs increases to cause the overall effective cost of operating LED lighting products to be lower than incandescent lamps and fluorescent lamps providing equivalent illumination. Also, LEDs can be dimmed by controlling the current through the LEDs because LEDs are current driven devices. The current through a plurality of LEDs in a lighting device must be controlled tightly in order to control the illumination provided by the LEDs. Typically, the secondary (output side) of an LED lighting device must be electrically isolated from the primary (line and neutral side) of the lighting device to meet applicable safety standards (e.g., IEC class II isolation). In addition, an LED driver circuit should desirably have a high power factor and constant current control.
Flyback-type DC-DC converters are conventionally known in the art for constant current supply based on their relative simplicity and the abundance of low-cost gate drive integrated circuits (IC) available in the market. Referring to FIG. 1, one example of an LED driver 100 utilizing an isolated flyback converter with constant output current control is shown. The LED driver 100 includes a primary side circuit 110, a secondary side circuit 130, and a feedback circuit 150. The primary side and secondary side circuits 110, 130 are electrically isolated via a flyback transformer T1. The transformer T1 includes a primary winding T1_P and a secondary winding T1_S.
The primary side circuit 110 includes the primary winding T1_P, a first switching element Q1, a gate drive integrated circuit (IC) 112, and a primary voltage source V_in. The primary voltage source V_in may be provided either from an input rectifier such as a diode bridge (not shown) or from a power factor correction circuit (not shown) output. Accordingly, the primary input voltage V_in is generally characterized herein as a DC voltage supply.
A conventional example of the gate drive IC 112 may be an L6562 controller from STMicroelectronics, and is configured to enable and disable gate drive signals 112_S to switching element Q1 based on various input signals. The gate drive IC 112 as simplified in FIG. 1 is coupled to the feedback circuit 150, and includes at least a gate drive logic circuit 114 and an error amplifier (OPAMP) 116. The gate drive OPAMP 116 includes at least an inverting input 116_I, a non-inverting input 116_NI, and an output 116_O. The output 116_O is configured to produce and transmit an error signal 118 to the gate drive logic circuit 114. The first switching element Q1 (e.g., a MOSFET) includes a drain node Q1_D, a gate node Q1_G, and a source node Q1_S. The gate node Q1_G is configured to receive the gate drive signals 112_S for enabling and disabling the first switching element Q1.
The primary side circuit 110 further includes a first resistor R1, a second resistor R2, a third resistor R3, a fourth resistor R4, a fifth resistor R5, a first capacitor C1, and a secondary voltage source Vcc. The primary winding T1_P is coupled between the primary voltage source V_in and the drain node Q1_D of the first switching element Q1. The fifth resistor R5 is coupled between the source node Q1_S of the first switching element Q1 and a primary side ground GND_P.
First and second resistors R1, R2 are configured as a divider network to generate and transmit a multiplier signal 120_M from a multiplier node 120 to the gate drive logic circuit 114. The first resistor R1 is coupled between the first node 120 and the primary voltage input V_in. The second resistor R2 is coupled between the first node 120 and the primary side ground GND_P.
The secondary voltage source Vcc is coupled between the primary side ground GND_P and voltage input 112_Vcc of the gate drive IC 112. The feedback circuit 150 is coupled to the inverting input 116_I of the gate drive OPAMP 116. Resistor R3 is coupled between the inverting input 116_I of the gate drive OPAMP 116 and the voltage input 112_Vcc of the gate drive IC 112. Resistor R4 is coupled between the primary side ground GND_P and the inverting input (INV) 116_I of the gate drive OPAMP 116. An integrating capacitor C1 is coupled between the inverting input 116_I of the gate drive OPAMP 116 and the output pin (COMP) 116_O of the gate drive OPAMP 116.
The non-inverting input 116_NI of the gate drive OPAMP 116 is configured to receive an internal reference voltage signal (V_ref) of the gate drive IC 112. The gate drive logic circuit 114 is configured to receive a switch current feedback signal 122 through the first switching element Q1 as sensed by resistor R5. The gate drive signals 112_S are provided to the gate node Q1_G of the first switching element Q1 via the gate drive logic circuit 114 based on signals from the feedback circuit 150 at the inverting input 116_I, the multiplier signal 120_M (via the first and second resistors R1, R2 and corresponding to the input voltage V_in), the switch current signal 122 (through the resistor R5), and the error signal 118 produced at the output 116_O of the gate drive OPAMP 116.
The secondary side circuit 130 includes the secondary winding T1_S, a diode D1, an output filter capacitor C2, and a load current sensing resistor R6 coupled in series with a load (R_load). The secondary winding T1_S is coupled between a secondary side ground GND_S and an anode of the diode D1. The output capacitor C2 is coupled between a cathode of the diode D1 and the secondary side ground GND_S. The diode D1 is configured to allow energy stored in the secondary winding T1_S to charge up the output capacitor C2 when the first switching element Q1 is turned off. An output node (e.g., between the cathode of diode D1 and a first terminal of the output capacitor C1) is also connected to a first terminal of the load, which may comprise, for example, one or more light-emitting diodes (LEDs) that emit light when sufficient current passes through the LEDs. A second terminal of the load is connected to a current sensing terminal 132 and to the first terminal of the current sensing resistor R6. A second terminal of the current sensing resistor is connected to the secondary circuit ground reference (GND_s). When current flows through the load, the same current flows through the current sensing resistor. Accordingly, a voltage develops on the current sensing terminal that has a magnitude with respect to the secondary circuit ground reference that is proportional to the current flowing through the load. In one embodiment, the current sensing resistor has a resistance of, for example, 0.1 ohm such that the effect of the resistance of the current sensing resistor on the load current is insignificant.
The feedback circuit 150 is coupled between the output current node 132 of the secondary side circuit 130 and the inverting input 116_I of the gate drive OPAMP 116. Because the intensity of the light emitted by the LEDs in the load (R_load) is dependent on the magnitude of the current flowing through the LEDs, the current is controlled closely. The current sensing resistor R6 senses the current going through the load and develops a voltage on the current sensing node proportional to the load current. The voltage representing the sensed current is fed back to a proportional integral (PI) current control loop to provide current regulation. In FIG. 1, the PI current control loop comprises an operational amplifier (OPAMP) 152 having an inverting (−) input terminal 152_I, a non-inverting (+) input terminal 152_NI, and an output on an output terminal 152_O. The current sensing node is connected to the inverting input of the operational amplifier via a series resistor R7. A feedback resistor R8 and a feedback capacitor C3 are connected in series between the output terminal of the operational amplifier and the inverting input. A reference current (I_ref) or a reference voltage having a magnitude corresponding therewith is connected to the non-inverting input of the operational amplifier. The reference current signal I_ref may for example be provided from an external dimming control device, a local user interface, one or more sensors, a lighting management system, or the like. The magnitude of the reference current may be selected to produce a desired load current through the load. The reference current may be a fixed reference current to provide a constant load current, or the reference current may be a variable reference current to allow the load current to be varied to thereby change the intensity of the light emitted by the LEDs in the load. For example, a reference voltage may be generated by a dimmer circuit (not shown) that selectively produces a plurality of voltage levels corresponding to a plurality of load currents, wherein each magnitude of load current corresponds to a light intensity. The operational amplifier is responsive to the relative magnitudes of the reference signal and the sensed output signal to provide feedback to the gate drive IC 112 as described below.
The output 152_O of the operational amplifier 152 is connected to a first input of a photocoupler 154. The photocoupler (also referred to as an opto-isolator or an optocoupler) has an internal light generation section (e.g., an LED) 154_D coupled to the input of the photocoupler. In the illustrated embodiment, the output voltage from the operational amplifier is applied to the cathode of the internal LED via the first input. The anode of the internal LED is connected via a second input of the photocoupler to a first terminal of a pullup resistor R9. A second terminal of the pullup resistor is connected to a secondary positive voltage source Vcc_s, which is referenced to the secondary ground reference GND_S. The voltage source also provides the supply voltage to the operational amplifier. The internal LED in the light generation section is responsive to a low voltage applied to the first input to generate light. The intensity of the generated light is responsive to the magnitude of the difference between the voltage on the first input and the secondary positive voltage. The generated light is propagated internally to the base of a phototransistor 154_E in an output section within the same component. The phototransistor is responsive to the generated light to vary the conductivity and thereby to effectively vary the impedance of the phototransistor. The phototransistor has a collector that is connected to a current control node between (previously described) resistors R3, R4. The phototransistor has an emitter that is connected to the primary circuit ground reference GND_P. As illustrated the photocoupler electrically isolates the secondary circuit voltages and the secondary circuit ground reference in the secondary circuit side 130 from the components in the primary circuit side 110.
When the voltage applied to the first input of the photocoupler 154 from the output 152_O of the operational amplifier 152 decreases (e.g., becomes more negative with respect to the secondary positive voltage (Vcc_s) connected to the pullup resistor R9), the conduction of the phototransistor 154_E in the output section of the photocoupler increases to effectively reduce the resistance from the current control node between resistors R3, R4 to the primary circuit ground reference GND_P such that the current from the feedback circuit 150 as a whole increases. The increased current increases the switching frequency of the gate drive IC 112.
When the voltage applied to the first input of the photocoupler 154 increases (e.g., becomes less negative with respect to the secondary positive voltage (VCC_S) connected to the pullup resistor R9), the conduction of the phototransistor 154_E in the output section of the photocoupler decreases to effectively increase the resistance from the control node between resistors R3, R4 to the primary circuit ground reference GND_P such that the current from the feedback circuit 150 decreases. The decreased current decreases the switching frequency of the gate drive IC 112.
The duty cycle of the on-time of the switching element Q1 decreases with increased frequency and increases with decreased frequency. A decrease in duty cycle causes the energy transferred to the load to decrease. An increase in duty cycle causes the energy transferred to the load to increase. Thus, the load current decreases with increased switching frequency, and the load current increases with decreased switching frequency.
From the foregoing, it can be seen that when the load current through the current sensing resistor R6 generates a voltage that is less than the voltage corresponding to the reference current I_ref, the output voltage of the operational amplifier 152 increases. The increased output voltage produced by the operational amplifier causes the photocoupler 154 to decrease the light generated between the input section and the output section, which causes the photoresistor 154_E in the output section to decrease conductivity and thus increase the effective impedance on the emitter 116_O of the gate drive OPAMP 116. The increased effective impedance decreases the voltage on the inverting pin (INV) 116_I of the OPAMP. The decreased voltage decreases the switching frequency of the gate drive IC 112, which increases the duty cycle of the switching voltage applied to the control input terminal of the switching element Q1. The increased duty cycle has the effect of increasing the energy transferred to the output filter capacitor C2, which increases the voltage on the output node, which increases the current flowing through the load (R_load). The load current will increase until the sensed load current is substantially equal to the reference current (I_ref).
When the current flowing through the load (R_load) is greater than the reference current (I_ref), the opposite transitions occur. The voltage on the output 152_O of the operational amplifier 152 decreases. The decreased output voltage produced by the operational amplifier causes the photocoupler 154 to increase the light generated between the input section and the output section, which causes the phototransistor 154_E in the output section to increase conductivity and thus decrease the effective impedance on the emitter 116_O of the gate drive OPAMP 116. The decreased effective impedance increases the voltage on the inverting pin (INV) 116_I of the OPAMP. The increased current increases the switching frequency of the gate drive IC 112, which decreases the duty cycle of the switching voltage applied to the control input terminal of the switching element Q1. The decreased duty cycle has the effect of decreasing the energy transferred to the output filter capacitor C2, which decreases the voltage on the output node, which decreases the current flowing through the load. The load current will decrease until the sensed load current is substantially equal to the reference current.
Critical mode (also known as boundary mode or critical conduction mode) gate drive IC's are conventionally known and popular at least in part for their relative simplicity. Referring next to FIG. 2, an exemplary operation of the aforementioned lighting device in a critical mode 200 may now be described. A primary current T1_PI through the primary winding T1_P of the transformer T1 starts from zero amperes and rises throughout application of gate drive signals 112_S and a corresponding on-time T_on of the switching element Q1, and a secondary current T1_SI through the secondary winding T1_S of the transformer T1 starts high when the gate drive signals 112_S are removed and falls during a corresponding off-time T_off of the switching element to zero amperes.
As shown in FIG. 2, the output current (I_out) through the load (R_load) is the average of the secondary current T1_SI in the secondary winding T1_S. The relationship between the primary current T1_PI and the secondary current T1_SI is as follows:T1_PI=N×T1_SI  (1)where N is the turns ratio between the primary winding and the secondary winding of the isolation transformer.
As shown in equation (1), a higher primary current T1_PI is associated with a higher secondary current T1_SI. A longer on-time T_on of the first switching element Q1 produces more current through the primary winding T1_P and correspondingly more current through the secondary winding T1_S, which will result in a higher output current I_out through the load R_load.
According to inductor voltage-time relationship, the on and off time T_on, T_off of the gate drive signal can be defined as:V_in×T_on=N≤V_load≤T_off=N×(I_out×R_load)×T_off  (2)where V_load is the load voltage and R_load is an equivalent resistance of the load at a certain output current I_out.
According to equation (2), when the output current I_out and the equivalent resistance of the load R_load reaches a certain value, the on-time T_on will be at a minimum on-time T_on_min and off-time T_off will be at maximum off-time T_off_max. For at least the purposes of the present disclosure, we may define this value as the start of discontinuous mode 300 (as shown in FIG. 3) operation having a corresponding discontinuous output voltage V_load_dis, a discontinuous output current I_out_dis, and a discontinuous equivalent resistance of the load R_load_dis. Equation (2) updated for the discontinuous mode of operation can be defined as:V_in×T_on_min=×V_load_dis≤T_off_max=N×(I_out_dis×R_load_dis)×T_off_max  (3)
By conventional design, a critical mode gate drive IC 112 has an internal minimum on-time T_on_min and a maximum off-time T_off_max. As such, if the output current I_out and equivalent resistance of the load R_load decreased further below the discontinuous output current I_out_dis and the discontinuous load resistance R_load_dis, then the gate drive IC 112 will not be able to further reduce its on-time T_on and increase its off-time T_off to compensate the output current I_out and/or load resistance R_load. As a result the gate drive IC 112 will be forced into a third mode of operation, defined for at least the purposes of the present disclosure as a random pulse mode.
As illustrated in FIG. 4, in the random pulse operating mode 400 when the equivalent load resistance R_load is very small (associated with a low output current and a low output voltage) the gate drive signals 112_S will randomly pulse to maintain a certain average output current I_avg of the load R_load. Since the gate drive signals 112_S are randomly pulsing, the output current I_out and load voltage V_load will be randomly changing, which will result in an annoying flickering of the lighting output from the load. This is highly undesirable at any time for LED lighting applications, so it is very important to keep the gate drive signals 112_S as consistent as possible to avoid LED flickering. It would therefore be desirable, for the purpose of avoiding flickering, that the gate drive IC remain in either the critical mode 200 or the discontinuous mode 300.
According to equation (3), when the output current I_out is less than the discontinuous output current I_out_dis and the equivalent resistance R_load of the load is less than the discontinuous equivalent resistance R_load_dis of the load, the gate drive IC would be able to maintain a regular continuous operating mode 200, 300 (to avoid the random pulsing mode 400) if the off-time T_off could be further increased to allow the gate drive IC entering into the discontinuous operating mode 300 to maintain that state of operation.
There are no available solutions for avoiding the random pulse mode when the gate drive integrated circuit is operated at its limits, or when the output current or equivalent resistance of the load are decreased below their discontinuous values.